The present invention relates to systems for receiving digital radio signals and, more specifically, the invention relates to receivers of radio signals encoded by a pseudonoise sequence, which are used in the global positioning systems GPS (Global Positioning System) (USA) and GLONASS (Global Navigational Satellite System)(Russia).
Global positioning systems such as GPS (Global Positioning System) (USA) and GLONASS Global Navigational Satellite System) allow a user with a passive receiver of digital signals to generate an exact definition of the user""s coordinates (longitude, latitude, altitude,) and time. (Cf. xe2x80x9cGlobal Navigational Satellite Systemxe2x80x94GLOSNASS. The Interface Control Document.xe2x80x9d KNITS VKS Russia, 1995. See also xe2x80x9cGlobal Position System. Standard Positioning Service. Signal Specificationxe2x80x9d. USA, 1993). The navigational radio signal transmitted by the global positioning system satellite is a multicomponent phase-manipulated signal, in which the signal of a carrier frequency L1 of about 1.6 GHz is modulated by a coherent pseudonoise binary sequence xc2x11 (phase manipulation on TL radian) having a length of 1023 characters (GPS) or 511 characters (GLOSSNAS). The pulse-repetition rate of the modulating sequence is equal to 1.023 MHz for the GPS and 0.511 MHz for the GLONASS, with the pulse-repetition period being 1 ms. Application of the method of digital reception and the correlation of a similar broadband digital signal allows for the successful reception and decoding of a very low amplitude signal located much below the level of natural thermal noise. Thus, in the case of the GPS C/A signal, its level is from xe2x88x92157 dBW up to xe2x88x92160 dBW so that at a standard density of thermal noise of xe2x88x92205.2 dBWILz and a minimum band of the radio-frequency channel of 2 MHz results in a signal-to-noise ratio of 14.8 dB to xe2x88x9217.8 dB.
In addition, the application of the method of reception and digital processing of broadband phase-manipulated signals allows one to reduce essentially the negative effect of the narrow-band interference which often results in a failure of reception of the narrow-band amplitude-modulated or frequency-modulated signals. Nevertheless, the suppression of the narrow-band (sinusoidal) interference for a digital receiver of a pseudonoise signal (PNS) is critical, especially in the case of high-power pseudonoise interference whose amplitude overcomes that of the thermal noise. Furthermore, the GLONASS is a system with a frequency division of the signals for the receiver based on the GLONASS system, while for the combined GPS/GLONASS receivers the width of the radio-frequency channel is broadened approximately to 10 MHz. The use of the xe2x80x9cnarrow correlatorxe2x80x9d technique also results in broadening the radio-frequency band of the receiver. (Cf. .J. Dierendonck, P. Fentor, N. Ford in Theory and Performance of Narrow Correlator Spacing in GPS Receiverxe2x80x9d, Navigation: Journal of the Institute of Navigation vol. xe2x88x9239, No.3, Fall 92). The extension of the range of the radio-frequency channel results in an increase of probability of catching a high-power narrow-band interference, and, as a consequence, it is necessary to provide means for dealing with this interference.
Known in the art is a method of using the adaptive analog-to-digital converters enabling the narrow-band interference on the digital PNS receiver operation to be reduced. (Cf. Frank Amoroso, Jacob L. Bricker xe2x80x9cPerformance of the Adaptive ADC in Combined CW and Gaussian Interference, IEEE Transactions and Communications, vol. COM-J4, No.3, March 1986)[1]. Using a two-bit adaptive ADC as a digitizer with a variable quantization threshold xcex94, it is possible to reduce significantly the effect of the narrow-band interference on the operation of a digital correlator.
Also known in the art is a receiver for decoding a complex signal consisting of many PNS. The receiver comprises a reference generator, an automatic gain control (AGC) device having an input for complex PNS and an input for a signal controlling the amplification factor, a multilevel adaptive ADC converter whose input is connected to the AGC output and the clock input is connected to output of the reference generator. The converter produces at its output in-phase I and quadrature Q components of the complex signal. The receiver also has a set of digital counters, in which each counter calculates a value of digitized signals in one of the channels whose amplitude is within a preset quantization interval, and a control device reading the output values of the counters and producing a gain control signal on the basis of the analysis of the obtained data. See Patric Fenton, Kkwok-Ki K. Ng,. Thomas J. Ford in xe2x80x9cMulichannel Digital Receiver for Global Positioning Systemxe2x80x9d, U.S. Pat. No. 5,101,416.
One of the feature of the present invention is that, given a multilevel ADC converter and calculating the percentage of the digitized signals appearing between two adjacent quantization thresholds, it is possible to evaluate how the distribution function of the digitized complex signal corresponds to the Gaussian. Thus, in an exemplary embodiment of the present invention, a 6-level complex quantizer is provided, at output of which the quadrature components can take values xc2x11, xc2x12, and xc2x13. By setting the value of the A-distances between quantization thresholds, one can achieve the necessary ratio of a value of digitized signals appearing in one or another quantization interval. In the present embodiment it is suggested to use a ratio of 49%, 32%, and 19% for the signals from the intervals +1, xc2x12, and xc2x13 corresponding to the quantization interval xcex94=.66"sgr", where a is the square root from the dispersion of the Gaussian distribution.
The deviation from the given distribution points to the presence of narrow-band interference which can be compensated for by changing the gain of the AGC circuit and the quantizer output values. In this embodiment the presence of the narrow-band interference is recorded, when all quantized values are only in four of the six quantized intervals, i.e. the distribution will be 49%, 51%, and 0% for the signals from the intervals of xc2x11, xc2x12, and xc2x13. In this case the presence of narrow-band interference stated and the quantization threshold is changed so that the digitized values in the intervals corresponding xc2x11 is equal to 85%; xc2x12 is xe2x88x9215%; and xc2x13 is xe2x88x920%. In so doing the numerical values of quantized values are changed: xc2x11 are replaced by 0, i.e. 85% of signals below the quantization thresholds xc2x1xcex94 are ignored; xc2x12 are replaced by xc2x11; and xc2x13 are replaced by 0. Thus, during further calculations only the approximately 15% of digitized signals occurring within xcex94  less than | signal amplitude | less than 2xcex94, are taken into account in correlator channels.
The disadvantages of the offered device include, first, the fact that it requires application of technically complex multilevel multibit analog-to-digital converters (AID), and, second, at some points of the distribution function it is possible to make only a rough estimate of the amplitude of the narrow-band interference, which, in turn, lead to a coarse gain control for changing the quantization thresholds. In the above example, the discrete adjustment: 49%, 51%, and 0% are replaced by 85%, 15%, and 0%. At each ratio Vsi/"sgr", where Asi is the sinusoidal interference amplitude and a is the Gaussian noise dispersion, it is possible to select the best value of the quantization thresholds to minimize the effect of the narrow-band interference on the useful signals.
The basic object of the invention is the development of a digital PNS receiver compensating the effect of the narrow-band interference and allowing the removal of the above-said disadvantages due to the direct detection of the narrow-band interference, estimation of its amplitude, and installation of an optimal quantization threshold for the measured ratio Vs/"sgr". The given solution also allows the ADC to be simplified to a maximum extent, because even the three-level quantizer (0, xc2x11) makes it possible to perform effective suppression of the interference signals by means of correctly selecting the quantization threshold value.
This result is achieved because the receiver is provided with a multichannel correlator, in which, in addition to the conventional correlation of the PNS channels, which are usually represented by a correlator with an exact copy of the pseudonoise signal and a correlator early-minus-late copies of the pseudonoise signal, it has an additional correlator for detecting the narrow-band sinusoidal interference. The given channel includes a digital controlled carrier generator, which generates quadrature phase counts. The channel also comprises digital correlators and accumulators for storing the quadrature components. The correlator also comprises digital counters counting the number of counts in each quantization interval for a predetermined time period. In the case of three-level quantization this number of counts equal to xe2x88x921, 0, or +1, calculated in an interval for providing statistical assurance, for example, such as N counts  greater than 104. By reading the values from the accumulators of the additional channel and comparing these values with the detection threshold, the processor makes a decision on the presence of narrow-band interference and evaluates its amplitude. Depending on the evaluated Vsi/"sgr" ratio, the processor determines the optimal relations between the quantity of counts for each quantization interval and adjusts the quantization thresholds xcex94 and amplification factor AGC. Thus, the value of thresholds xcex94 can be checked by the amount of counts of the value being quantized within the preset quantization interval.
Generally, the essence of the described invention is that when a sinusoidal noise signal is superimposed on the PNS noise, the signal-to-noise after the quantization depends essentially on the sinusoidal interference and on the quantization thresholds. This dependence is illustrated by FIG. 1, where in graph 1(a) the a pure pseudonoise signal is shown, and in graph 1(b) illustrating the same signal but in a combination with the sinusoidal interference. As it is clearly seen in graph 1(b), the number of the PNS chips which can be distinguished on the interference background, increases at a choice of a threshold V2 greater than V1. Also, setting the digitization value in such a manner that the signal at the output of the ADC A =0 corresponds to all V of the  less than V2 signal, removes the effect of the noise caused by the ambiguity of the PNS resolution on the background of the high-power sinusoidal interference.
The problem of selection of the useful PNS on a noisy background is much more difficult, as the useful signal is received on the background of a combination of the thermal Gaussian noise and sinusoidal interference. Thus, the distribution of the probability density of the signal value in the case of sinusoidal interference Vosin((xcexa9t) is described by the function:                     ρ        ⁡                  (          v          )                    =              1                  π          ⁢                                    1              -                                                (                                      v                    /                                          v                      0                                                        )                                2                                                          ,                  "LeftBracketingBar"        v        "RightBracketingBar"            ≤              v                  0          ⁢                      
                                                  ρ        ⁡                  (          v          )                    =      0        ,                  "LeftBracketingBar"        v        "RightBracketingBar"             greater than               v        0            
Using a method stated in [1], and considering that Gaussian thermal interference, sinusoidal interference, and the digital signal are values that are statistically independent, the efficiency of digitization of a signal through an effective amplification factor can be determined:       G    =                  [                                            V              a                        /                          σ              a                                                          V              i                        /                          σ              i                                      ]            2        ,
where Vi, "sgr"i Va, "sgr" are the amplitude and dispersion of the signal before and after the ADC, "sgr"i2=N+I; N is the Gaussian noise power; and I is the sinusoidal interference power.
The Va,"sgr"a for the three-level analog-to-digital converter can be calculated from the following calculations
            P      ⁡              (                  Va          =                      +            1                          )              =                  ∫                  Δ          -                      V            i                          ∞            ⁢                        H          ⁡                      (            y            )                          ⁢                  xe2x80x83                ⁢                  ⅆ          y                      ,      
    ⁢            P      ⁡              (                  Va          =                      -            1                          )              =                  ∫        ∞                  Δ          -                      V            i                              ⁢                        H          ⁡                      (            y            )                          ⁢                  xe2x80x83                ⁢                  ⅆ          y                      ,      
    ⁢            P      ⁡              (                  Va          =          0                )              =                  ∫                  Δ          -                      V            i                                    Δ          -                      V            i                              ⁢                        H          ⁡                      (            y            )                          ⁢                  xe2x80x83                ⁢                  ⅆ          y                      ,
where (Va=+1); (Va=xe2x88x921); and (Va=9) are the probability of generation of values +1, xe2x88x921, 0 at the output of the ADC with the presence of the input signal Vi and quantization threshold xcex94. The distribution density of input signal amplitude H(y) is determined from the common probability distribution of the amplitude of the Gaussian and sinusoidal interference:                     H        ⁡                  (          y          )                    =                        ∫                      -            ∞                                +            ∞                          ⁢                              ρ            ⁡                          (              v              )                                ⁢                      G            ⁡                          (                              y                -                v                            )                                ⁢                      xe2x80x83                    ⁢                      ⅆ            v                                ;                      ρ        ⁡                  (          v          )                    =              1                  π          ⁢                                    1              -                                                (                                      V                    /                                          V                      0                                                        )                                2                                                          ,                  "LeftBracketingBar"        V        "RightBracketingBar"            ≤              V                  0          ⁢                      
                              xe2x80x83xcfx81(v)=0, |V| greater than V0
is the sine-wave signal distribution; and             G      ⁡              (        v        )              =                  1                              2            ⁢                          πσ              2                                          ⁢              e                                            -                              v                2                                      /            2                    ⁢                      σ            2                                ,
is the Gaussian thermal noise distribution characterized completely by the dispersion "sgr". For numerical calculations the values H(y) and P(Va) as the infinite limits of integration are replaced by values, where the Gaussian distributions function may be considered equal to zero. Usually, the value 3"sgr" is sufficient. Thus, the formulas of calculations will take a form
                    P        ⁡                  (                      Va            =            1                    )                    =                        ∫                      Δ            -                          V              i                                                          V              0                        +                          3              ⁢              σ                                      ⁢                              H            ⁡                          (              y              )                                ⁢                      xe2x80x83                    ⁢                      ⅆ            y                                ;    and                      P        ⁡                  (                      Va            =                          -              1                                )                    =                        ∫                                    -                              V                0                                      -                          3              ⁢              σ                                                          -              Δ                        -                          V              i                                      ⁢                              H            ⁡                          (              y              )                                ⁢                      xe2x80x83                    ⁢                      ⅆ            y                                ;  
Where V0 is the amplitude of sinusoidal interference.
The value V, is calculated proceeding from the values of probability P:
E(Va)=Va+(xe2x88x921)*P(Va=xe2x88x921)+(+1)*P(Va=+1)
Providing allowance for the smallness of Va:
"sgr"axe2x89xa1E(V2a)=P(Va=xe2x88x921)+P(Va=+1).
Thus, it is possible to calculate the effective gain G knowing the ratio between the amplitudes of the signal being detected, sinusoidal interference signal and Gaussian noise dispersion.
The analysis of the dependence of the optimal value G on xcex94 and Vi has shown that when Vi  less than  less than V0, V0 less than  less than "sgr", G very poorly depends on Vi. Depending on the ratio of the sinuisoidal interference amplitude to the noise dispersion V0/"sgr", we may offer the following simplified choice of installation of the thresholds xcex94:
At V0xe2x89xa60.5"sgr", the amount of 0 th counts =50%, the amount of +1=50%; at 0.5"sgr" less than V0 less than 2"sgr", the amount of 0 th counts =70%, that of xc2x1130%; at V0 greater than 2"sgr" the amount of 0 th counts =85%, the amount of xc2x11=15%.
The above analysis has been made for the case of the three-level quantization, however, it can easily be extended for ADC with any number of the levels.
It should also be noted that the narrow-band interference detector may be made in the form of a tracking channel correlator with disconnected code generator, i.e. a code generator should simply generate 1 instead of a pseudonoise sequence.